RF telemetry receiver circuit for active medical implants

ABSTRACT

An RF telemetry receiver circuit for active implantable medical devices. The baseband binary signal (D b ) is doubly modulated by a low frequency carrier (F m ) and by a high frequency carrier (F c ). The receiver circuit is a semi-passive non heterodyne circuit, devoid of a local oscillator and mixer. It comprises an antenna ( 104 ), a passive bandpass filter ( 108 ) centered on the high-frequency carrier (F c ), a passive envelope detector ( 120 - 126 ) and a, digital demodulator ( 116 ). The envelope detector comprises a first diode circuit ( 120 ) of non-coherent detection, an active bandpass filter ( 122 ) centered on a frequency (2.F m ) twice the low frequency carrier and having a bandwidth (2.D b ) twice the baseband bandwidth, and a second diode circuit ( 124 ) of non-coherent detection, outputting a baseband signal applied to the digital demodulation stage ( 116 ).

RELATED APPLICATIONS

The present application claims the priority date benefit of FrenchApplication No. 11/55759 entitled “RF telemetry receiver circuit foractive medical implants” and filed Jun. 28, 2011, which is herebyincorporated by reference in its entirety.

FIELD

The present invention relates to “medical devices” as defined by theJun., 14, 1993 Directive 93/42/CE of the European Community Council, andmore particularly to “active implantable medical devices” as defined bythe Jun. 20, 1990 Directive 90/395/CEE of the European CommunityCouncil. Such devices include in particular apparatus responsible formonitoring cardiac activity and generating stimulation pulses forresynchronization, defibrillation and/or cardioversion, in case of anarrhythmia detected by the device. It also includes neurologicaldevices, pumps for the diffusion of medical substances, cochlearimplants, implanted biological sensors, etc., and devices for measuringpH or intracorporeal impedance (such as transpulmonary impedance orintracardiac impedance). It should be understood that the presentinvention is particularly advantageous with implanted devices such aspacemakers, cardioverters or defibrillators, and equally withnon-implanted medical devices, such as data recorders like externalHolter devices intended for ambulatory patient monitoring and recordingof physiological parameters such as cardiac activity.

BACKGROUND

Active medical devices are generally designed to enable a bidirectionaldata exchange with a remote “programmer,” which is an external device,for a variety of reasons, e.g., to verify the configuration of thedevice, to read information the device has recorded, to post informationto the device, or to update the internal software of the device.

Techniques for monitoring patients at home (home monitoring) have beenproposed that implement a remote interrogation device disposed adjacentto the implanted patient, which is periodically activated, e.g., daily,to download data collected by an implanted device and transmission foranalysis to a remote surveillance site. Data exchange between animplantable device and the programmer or home monitoring device(hereinafter, collectively an “external device”) is generally performedby telemetry, that is to say, by a communication technique for remotetransmission of information, without galvanic contact.

A common telemetry technique uses inductive coupling between coilslocated inside the implanted device and those in the external device,which technique is known as “induction telemetry”. The inductiontelemetry technique has the disadvantage, because of the very limitedrange of such a coupling, to require the use of a “telemetry head”connected to the external device and containing a coil that an operatorplaces in the vicinity of the site in which the device is implanted.

It was recently proposed to implement an alternative technique fornon-galvanic coupling, using an electromagnetic wave generated by andbetween complementary transmitter/receiver circuits operating in theradiofrequency (RF) domain, typically in the range of frequencies ofseveral hundred megahertz. This technique, known as “RF telemetry”allows programming or interrogating implants at distances greater than 3m, and therefore permits the exchange of information withoutmanipulation of a telemetry head, and indeed even without externaloperator intervention.

An active medical device implementing RF telemetry is described forexample in EP1862195A1 and its US counterpart U.S. Pat. No. 7,663,451(both assigned to Sorin CRM S.A.S., previously known as ELA Medical).

Power Consumption by RF telementary circuits is a crucial aspect of thisRF technique, especially with regard to an implanted device. Indeed, theimplanted devices use for energy sources a battery whose energy capacityis limited (about 1 Ah). Without RF telemetry, the average consumptionof an implanted device is about 20 μW, which provides autonomy (i.e., auseful life under normal circumstances) of about a dozen years. However,upon activation of a RF telemetry function, an implanted devicetypically consumes, with current technologies, a dozen milliwatts, about500 to 1000 times more than its average consumption for the usualfunctions of cardiac sensing and pacing.

On the other hand, it is known that the environment surrounding thepatient has more and more radio frequency interferences that couldinadvertently and adversely wake up the RF telemetry circuit of theimplanted device and therefore unnecessarily discharge the battery ofthe implanted device.

Occasional use of the RF telemetry once every three or six months, forexample, in connection with follow-up visits to the cardiologist, doesnot significantly degrade the autonomy of the implanted device. However,daily use to communicate with a home monitoring device can significantlyreduce the lifetime of the implanted device. Indeed, if for example aquarter of the capacity of the battery is assigned to the RF telemetryfunctions, this capacity provides only 50 hours of continuous RFcommunication throughout the useful life of the implanted device (10years), or a daily use of less than 50 seconds.

Knowing that the current RF telemetry systems use communication channelsof bandwidth limited to 300 kHz with a data rate of about 100-200 kbps(kilobits per second), and that the electrogram (“EGM”) signals aresampled on 10 bits and 500 times per second, a transmission duration of50 seconds per day allows transmission to the external device (takinginto account the overload introduced by the communication protocol) ofonly a maximum of 8 minutes recording of two EGM signal channels.

There is therefore a need for an intensive use of a remote RF telemetrysystem that is efficient in terms of energy use without penalizing theautonomy of the implanted device.

OBJECT AND SUMMARY

The starting point of the present invention lies in the implementationin the implanted device of communication circuits whose powerconsumption and complexity are minimal, even at the expense ofincreasing the complexity of circuits and power consumption of theexternal devices. Indeed, the latter operates with a not limited powersource available from a commercial line service or a battery of a muchlarger capacity than the battery of an implanted device and are alsoeasily replaceable or rechargeable.

In accordance with the present invention, the RF signals originatingfrom the external device are chosen with a modulation structure to makethe reception circuit in the implanted device as simple as possible andof very low power consumption, such that the complexity and theadditional consumption to generate this specific RF signal modulation ismanaged by the external device.

Broadly, the present invention is directed to an RF telemetry receptioncircuit that is in itself known, for example, according to the documentDE 196 46 746 A1 and its counterpart U.S. Pat. No. 6,453,200 B1, whichis incorporated herein by reference, for active medical implantabledevices, an RF signal which has a binary signal at a baseband data ratethat contains the information to be transmitted, which undergoes adouble-modulation for the transmission by RF telemetry, in which thebinary base band signal undergoes a first modulation by a first carrierfrequency to produce a first modulated signal, and that first modulatedsignal is then subjected to a second modulation by a second carrierfrequency, thereby provided the double modulated signal to betransmitted. The first carrier frequency is a lower frequency carrierthan the second carrier frequency.

The reception circuit successively comprising: an antenna for receptionof an RF signal, here the double modulated RF signal; a band-passfiltering and impedance matching stage that operates on the received RFsignal captured by the antenna; a envelope detector stage that receivesand operates on the filtered signal delivered by the front filteringstage and performs a first demodulation of the received and filtered RFsignal, and a digital demodulation stage that receives and operates onthe first demodulated RF signal delivered by the envelope detector stageand produces the binary signal at the base band containing theinformation to be transmitted.

The front filtering stage is centered on the frequency of the second orhigh-frequency carrier. The envelope detector stage is a non heterodynedetection stage, without any local oscillator or mixer, and comprises: afirst non-coherent detection diode circuit, receiving as input thefiltered signal delivered by the upstream filtering stage; an activebandpass filter, receiving as input the signal delivered by the firstnon-coherent detection diode circuit, this active band-pass filter beingcentered on a frequency twice that of the first or low frequency carrierand having a frequency bandwidth that is twice that of the base band,and a second diode non-coherent detection circuit, receiving as inputthe signal delivered by the active bandpass filter and outputting abaseband signal applied to the digital demodulation stage.

Most advantageously, the optimization also focuses on the choice of thefrequency bands used for transmission of RF signals. Moreover, bothregarding the modulation mode and the available channels in the selectedfrequency band, the present invention implements a non-symmetricalconfiguration between the two communication directions (a “Downlink”configuration from the external device to the implanted devices and“Uplink” configuration from the implanted device to the externaldevice).

The present invention also advantageously results in the Downlinkconfiguration reducing the consumption of the RF telemetry system by oneto two orders of magnitude (a improvement factor of from ×10 to ×100)compared to prior known. RF telemetry systems, thus maintaining a verylong autonomy of the implanted device even with intensive use of the RFtelemetry information exchange.

The present invention advantageously results in a modulation structureof the RF signal transmitted from the external device to the implanteddevice, whereby the Downlink configuration direction can greatly reduceRF interference, thereby avoiding degradation of energy consumption andtherefore the autonomy of the implanted device by an inadvertent oradverse wake up of the RF telemetry reception circuit.

In one embodiment, the reception circuit of the present invention is a“semi-passive” circuit, meaning that the components that are energyconsumers are reduced to a minimum number. In particular, this circuitneither includes a local oscillator or a mixer, unlike conventionalheterodyne demodulator reception circuits. The reception circuit caneven be devoid of a Low Noise Amplifier (“LNA”).

More particularly, this reception circuit is not only “semi-passive” butalso particularly selective, making it usable in particular frequencybands shared by many users, allowing the choice of a band that may bemore appropriate than that usually devoted to the RF transmission in thefield of medical implanted devices.

In one embodiment, the first diode non-coherent detection circuitcomprises a diode bias current source:

In one embodiment, the first diode non-coherent detection circuitcomprises a two diode detector configured as a voltage doubler.

In one embodiment, the frequency of the high-frequency carrier is in the2.4 GHz ISM band or in the 900 MHz RFID band.

In one embodiment, the frequency of the low frequency carrier is between25 and 500 kHz.

In one embodiment, the frequency of the baseband signal is between 5 and100 kbps.

In one embodiment, the gain of the active bandpass filter is between 30and 50 dB.

BRIEF DESCRIPTION OF THE DRAWINGS

Further features, characteristics and advantages of the presentinvention will become apparent to a person of ordinary skill in the artfrom the following detailed description of preferred embodiments of thepresent invention, made with reference to the drawings annexed, in whichlike reference characters refer to like elements, and in which:

FIG. 1 is a schematic view of the functional blocks involved in the RFtelemetry data exchange for an implanted device and an external devicein accordance with the present invention;

FIG. 2 is a schematic representation of the main functional blocks ofthe emission portion of the external device of FIG. 1.

FIG. 3 schematically illustrates the main functions implemented in aheterodyne reception circuit of the prior art;

FIG. 4 schematically illustrates the main functions implemented in aSchottky diode non-coherent detection reception circuit according to theprior art;

FIG. 5 shows a series of signal timing diagrams in various parts of theexternal device transmitter circuit shown in FIG. 3;

FIG. 6 is homologous to FIG. 5, showing the distribution of the varioussignals in question, considered in the frequency domain;

FIG. 7 schematically illustrates the main functions implemented by thesemi-passive reception circuit of the implanted device of FIG. 1,according to the present invention;

FIG. 8 shows a variant of the circuit of FIG. 7, with bias current andvoltage doubler; and

FIG. 9 shows a series of timing diagrams of signals recorded at variouspoints in the reception circuit of FIG. 7.

DETAILED DESCRIPTION

With reference to the drawing FIGS. 1-9, an implementation of thepresent invention in accordance with preferred embodiments will now bedescribed. With reference to FIG. 1 the circuits of an RFemission/reception system of an external device 10 communicating with animplanted device 100 are schematically illustrated. External device 10is equipped with a modulator (“MOD”) 12 delivering a signal to a poweramplifier 14 powering an emission antenna 16. For the reception partinside the external device, the signals collected by the antenna 16 areamplified by a low noise amplifier (LNA) 18 powering a demodulator(“DEM”) 20.

Implanted device 100 comprises a modulator circuit (MOD) 102 powering anantenna 104 for data transmission to external device 10 (i.e, an RFtransmission in the uplink direction). In the downlink direction (i.e.,the direction from external device 10 to implanted device 100), ademodulator (DEM) 106 collects the signals received by antenna 104 ofimplanted device 100.

The RF telemetry data exchange between external device 10 and implanteddevice 100 may be asymmetrical. More specifically:

-   -   The modulation types are not the same in both directions (that        is to say that modulators 12 and 102 do not operate using the        same modulation technique, and therefor as well demodulators 20        and 106, consequently), and    -   In the selected frequency band, different channels can be used        in one direction and in the other.

With reference to FIG. 2, modulation stage 12 of the external device 10is illustrated. It includes a local oscillator 22 the frequency F_(c) ofwhich is applied to a mixer 24 also receiving the digital signal to betransmitted (logic signal SL). The resulting signal, after passingthrough power amplifier 14 is applied to antenna 16 via a selectivebandpass filter 26 to limit the residual spectrum of the signal outsidethe considered transmission band.

FIG. 3 schematically illustrates the main functions implemented in aheterodyne reception circuit according to the state of the art, for thereception and the demodulation of the signal emitted by the externaldevice 10.

In such a conventional receiver (Cf. the Wikipedia article “Récepteurradio”), the signal collected by antenna 104 is applied to an impedancematching and band-pass filtering circuit 108, to maximize thesignal/noise ratio at the input of the reception circuit. The resultingsignal is applied to a low noise amplifier (LNA) 110 which amplifies thereceived signal in the frequency band of the carrier F_(c). Thedemodulation is carried out by a local oscillator 112, the frequency ofwhich is F_(c) and a mixer 114 receiving the filtered received signal.The baseband signal obtained by frequency translation is then applied toa digital demodulation stage 116 to restore the output logic signaltransmitted from the original external device.

It should be understood that this conventional structure of thereception circuit includes amplifier 110, local oscillator 112 and mixer114 as active circuits, which can represent up to 90% of the overallconsumption of the reception circuit (the digital demodulation stage116, which operates at low frequency, has a relatively low powerconsumption).

FIG. 4 shows an alternative reception circuit, also of the prior art(Cf. for example U.S. Pat. No. 5,929,779 A, FIG. 3), having no localoscillator and no mixer stage. These elements are replaced by a diode118 for non-coherent detection, for example, a Schottky diode, which isa passive component. The main active component of this embodiment is anenergy amplifier 110, which leads to consumption two to three times lessthan that of the conventional heterodyne detection circuit of FIG. 3.This reception circuit has a drawback due to the fact that the diodedetector 118 operates according to a quadratic voltage conversion of theamplified RF signal. It can be shown that such a receiver is therefore anon-selective receiver, which therefore simultaneously senses allpresent RF signals in the communication band defined by filter 108. Thislack of selectivity makes this type of receiver unusable in frequencybands shared by many users. It is certainly possible to improve theirselectivity by very narrow band filters, but they are difficult toimplement in an implanted device because of their large size.

In any event, even if such a receiver is tuned on a single channel ofcommunication to eliminate interference from other channels of the sameband of communication, the energy consumption of the amplifier 110remains high (a few milliwatts) because it operates in High Frequencyand does not allow extensive use of this receiver in a cardiac implanteddevice whose permanent consumption is, as noted above, only a few tensof microwatts.

The technique used by the present invention to overcome these variousdisadvantages of the prior art will now be described.

The present invention uses a double modulation of the signal in thedownlink direction (from the external device to the implanted device),and the external device is adapted accordingly. More precisely, withreference to FIG. 2 described above, in addition to the modulation bythe main carrier at frequency F_(c) (oscillator 22 and mixer 24), aprior modulation of the signal is expected by a lower frequency F_(m),by another local oscillator 28 associated with another mixer 30.

FIG. 5 illustrates the timing diagrams of the signals recorded atdifferent points a to e of the modulation circuit of the external deviceof FIG. 2. The timing diagram a illustrates the logic signal SLcontaining information to be transmitted. This signal is produced at adigital rate D_(b) (baseband signal) on the order of 5 to 100 kbps. Onecan for example use a NRZ (non return to zero) modulation in which thebinary ones are represented by a logic signal of given amplitude V, andlogic zeros by zero amplitude.

This signal a in the baseband is subjected to a double modulation,respectively by the F_(m) low frequency carrier signals and then by highfrequency carrier signals F_(c). It is specified here that the terms“low frequency” and “high frequency” are relative terms (the twofrequencies are different, and one is higher than the other) and do notimply any particular connotation on the chosen value, in absolute termsfor each of these frequencies. The numerical values that are givenshould be considered only as illustrative values, without limitation.

The low frequency carrier F_(m) (signal b) is advantageously at afrequency of the order of five to ten times the frequency of thebaseband Db, that is to say a frequency between 25 and 500 kHz.

The high frequency carrier F_(c) (signal c) corresponds to the selectedRF band, which is advantageously the 2.4 GHz band (more precisely, theband between 2.40 and 2.45 GHz), which is the unlicensed public ISM band(Industrial, Scientific and Medical). It could also be any authorized“RFID” band in the MHz range.

This 2.4 GHz ISM band is indeed preferred to the MICS band (MedicalImplant Communication System) 402-405 MHz generally used by medicaldevices, for the following reasons. In the 2.4 GHz band, an RF signaltransmitted from an external device located two meters from the patientto an implanted device located a few centimeters under the patient'sskin is attenuated in the order of 70-80 dB. In the MICS band, themaximum permissible power is only −16 dBm (25 mW). However, in the ISMband, the maximum permitted power is 20 dBm (100 mW), and can reach 30dBm (1 W) if one uses digital modulation techniques such as FrequencyHopping. It is thus possible to have a much higher power—on the order of1000 to 10 000 times higher (30 to 40 dB)—and so obtain more power intothe implanted device, with a lesser signal amplification. Thus, for asignal whose emitted RF power is between 10 dBm and 30 dBm (10 mW to 1W) and an attenuation of 70 to 80 dB, the power of the RF signalreceived by the implanted device, before amplification, is comprisedbetween −40 dBm and −70 dBm (100 mW and 100 pW).

The timing diagram d represents the logic signal after modulation by thelow frequency carrier (product of signals a and b), and the timingsignal e represents the same signal after the second modulation by thehigh-frequency carrier F_(c) (product of signals d and c).

FIG. 6 shows the distribution of the different signals a to a in thefrequency domain. The spectrum of the signal a is that of the basebandbinary signal D_(b), centered around the zero frequency, and occupiesthe frequency band between −D_(b) and +D_(b). The spectrum of the signalb is that of the low frequency carrier, and includes a peak at thefrequency +F_(m) and a peak at the frequency −F_(m). The spectrum of thesignal c is that of the high frequency carrier, and includes a peak atthe frequency +F_(c) and a peak at the frequency −F_(c). The spectrum ofthe signal d (after the first modulation) comprises the spectrum of thebaseband signal a, centered on the frequency +F_(m). The occupiedbandwidth is 2 D_(b). Finally, the spectrum of the signal e (after thesecond modulation) contains twice the spectrum of the baseband signal,with a sub-band centered around F_(c)−F_(m), and another aroundF_(c)+F_(m), both sub-bands therefore being separated by 2.F_(m).

The double modulation is certainly not spectrally efficient, because onecan show that the spectrum of the transmitted signal occupies four timesthe bandwidth of the baseband binary signal. However, although thisdouble modulation is not optimal from a spectral point of view, it canbe seen later that it permits to realize a semi-passive selectivereceiver in the implanted device which is advantageous for otherreasons.

This selectivity, very important to get a good quality receiver,exploits the fact that the two sub-bands of the spectrum of thetransmitted signal (signal e) are always separated by 2.F_(m) regardlessof the frequency of the high frequency carrier F_(c).

The semi-passive reception circuit of the implanted device, according tothe present invention, will now be described with reference to FIGS. 7and 9. This reception circuit is specifically adapted to utilize thedouble modulation technique as described above. The doubly modulated RFsignal is received by antenna 104. This signal is applied to animpedance matching and passive bandpass filtering stage 108 centered onthe frequency of the high-frequency carrier F_(c) to attenuate signalsoutside the selected band. The signal thus filtered is then applied to afirst non-coherent detection circuit 120, comprising a diode, preferablya Schottky diode, which performs the envelope detection by rectifyingthe signal according to a square (quadratic) equation.

The signal delivered by the first envelope detector 120 (signal f inFIG. 9) is then applied to a filter 122 which, typically, is an activebandpass filter centered around the 2.F_(m) frequency, and having abandwidth of the order of 2.D_(b). Such an active filter 122 is easy toimplement in standard CMOS technology, which consumes little energy.Filter 122 has a gain high enough to make detection of the signal in thefollowing stages of the reception circuit feasible. Preferably, it ispossible to produce such an active filter 122 with a voltage gain of 100(40 dB) with a consumption of about 10 μA.

It should be understood that the double modulation (low frequencycarrier and high-frequency carrier) used by the circuit of the presentinvention overcomes the limitations of a conventional non-coherentdetection circuit with diode of the prior art such as the one shown inFIG. 4. Indeed, in such a detector operating on a single carrierfrequency, we get only at the output of the detection stage a DCcomponent and this is for all frequencies within the useful band, thatis to say the band selected by the upstream filter 108. This DCcomponent is very noisy and corrupted, resulting in a poor transmissionquality of the digital signal from the external device to the implant.

The double modulation used by the present invention, however, allows amuch more robust transmission because, due to the double modulation,detection by diode 120 generates at its output not a DC component, butinstead a signal corresponding to the frequency of bandpass filter 122.Thus, the downstream signal processing for extracting the digitalinformation is made on the basis of the component located around thefrequency 2.F _(m), and not on a noisy and corrupted DC component.

It also should be understood that the signal f at the output of thefirst envelope detector circuit 120 after filtering of the highfrequency component by filter 108 has a signal/noise ratio of +5 dB,well below the tangential sensitivity of +8 dB of the diode of circuit120. Indeed, although the signals at the output from a diode detectorare very small, the noise level in these very weak signals determinesthe receiver sensitivity. In this case, in view of the double modulationof the signal, it is possible to detect a very low signal level at theoutput of diode detector 120. The sensitivity of this detector can becharacterized by the tangential sensitivity parameter (“TSS”) of thesignal, which is defined as the signal level which gives a signal/noiseratio equal to 8 dB at the output of the detector.

This tangential sensitivity is related to the bandwidth of the amplifierstage located after (and not before) of the detector according to theequation TSS=K√B, K being a constant and B being the bandwidth at theoutput. This relationship shows that the sensitivity of the diodereceiver can be greatly improved by reducing the bandwidth of the filterlocated at the output of the detector, i.e. by increasing theintegration time of the output signal.

In accordance with a preferred embodiment of the invention, theamplifier of active filter 122 has a bandwidth B=2.D_(b), as indicatedabove. The constant K between the bandwidth B and the TSS is on theorder of 1.4×10⁻¹² in the diode detector of the HSMS-285X family ofAvago Technologies, Inc. For a data rate of 5 kbps, this bandwidth is 10kHz, a TSS sensitivity of −68.5 dBm; for a data rate of 100 kbps, theTSS is equal to −62 dBm.

The timing diagram g of FIG. 9 shows the signal obtained at output ofactive bandpass filter 122. This signal is applied to a second diodeenvelope detector circuit 124 outputting a signal (h in FIG. 9)corresponding to the absolute (or rectified) value of the inputtedsignal g. This circuit is followed by a low-pass filter 126 fitted to acutoff frequency corresponding to the baseband frequency D_(b). Theresulting signal is illustrated in i in FIG. 9.

The latter signal is applied to a digital demodulation stage 116, forexample constituted by a hysteresis comparator combined with a counter,which retrieves at the output (signal j) the original signal deliveredby the external device (signal a) with simply a time offsetcorresponding to the integration time of the circuits. The counter canbe operated with a low-power clock at low frequency, of the order of5.D_(b) that is to say of the order of 25 to 500 kHz. Consumption ofsuch a digital demodulation circuit 116 may be less than 1 μA.

It should be understood that the reception circuit according to thepresent invention as described above does not contain any active circuitoperating in the frequency band of the high-frequency carrier F_(c),thus limiting the consumption of the reception circuit.

The total consumption of the reception circuit illustrated in FIG. 7 maythus be well below 15 μA, which is about 100 to 500 times less than thebest heterodyne receivers available today.

As regards the sensitivity of the receiving circuit, it is essentially afunction of signal/noise ratio at the stage containing the firstenvelope detector 120.

It is generally believed that the sensitivity of a receiver fornon-coherent detection (such as that shown in FIG. 4) is much lower thanthat of a heterodyne receiver (such as that shown in FIG. 3): thus, thenon coherent envelope detection receivers of in the ISM band 2.4 MHzgenerally have sensitivity between −40 and −50 dBm.

This would suggest that such an assembly would be difficult to use in animplanted device, wherein the power of received signals typically variesbetween −40 and −70 dBm, depending on the emission power. The circuit ofthe present invention, however, operates precisely contrary to thisgenerally accepted idea.

It is possible to further improve the sensitivity of the receptioncircuit as described above. In particular, in a preferred embodiment,providing a high input impedance for the amplifier stage of activebandpass filter 122 improves sensitivity and allows the use of alow-frequency carrier at a relatively low F_(m) frequency.

Another method to improve sensitivity by reducing consumption is to usefor stage 120 two diodes 130, 132 connected in a voltage doubler, asshown in FIG. 8. The use of two diodes to double the voltage increasesthe level of the input signal of active filter 122, and thereby reducesthe gain of this amplifier, with a corresponding reduction in powerconsumption.

Yet another method to increase the sensitivity is, as shown also in FIG.8, to inject into the diode(s) a bias current on the order of 100 nA to10 μA by means of a current generator 128. The value of this biascurrent is chosen to maximize sensitivity depending on thecharacteristics of the diode used, taking particular account of the factthat the tangential sensitivity TSS is not linear. Bias current, fixedor variable as a function of the received power, thus optimizes theconversion gain energy/voltage of the diode(s).

Compared with a circuit such as that shown in FIG. 7 without biascurrent and with a single diode, the circuit configuration illustratedin FIG. 8 provides a gain of 6 dB and a sensitivity of about −75 dBm fora data rate D_(b) of 5 kbps. Despite this sensitivity of −75 dBm beingless than that obtained with a heterodyne receiver (on the order of −100dBm for a data rate of 100 kbps), it is largely sufficient to obtain avery good receiver for low signal data rates. Indeed, in the downlinkdirection (from the external device to the implanted device) the volumeof information to be transmitted is relatively limited, since it mainlyconsists of commands and parameters sent to the implant—unlike theuplink direction, wherein a large amount of data stored in the memoryhas to be downloaded from the implanted device, such as EGM datacollected over a period of 24 hours or even days.

Moreover, in terms of energy consumption, the 10 mW consumption of aconventional heterodyne receiver for a data rate of 200 kbps leads to acost of 50 nJ/bit, while the receiver of the present invention consumesless than 50 μW for a data rate of 100 kbps, an energy cost of 0.5nJ/bit, a hundred times less.

In addition to reducing consumption, simplicity of construction of thereception circuit (essentially, two diode detectors and a low frequencyselective amplifier) allows a very simple hardware implementation,without any active component operating in the band of RF frequencies.

Yet another advantage of the receiver according to the present inventionis its ability to take into account a signal whose RF carrier frequency(the high frequency carrier F_(c)) varies in time, for example, in caseof modulation with frequency hopping to secure the reception circuitonto search among several possible channels one that is less noisyand/or provides the best transmission: indeed, the first envelopedetection circuit 120 is insensitive to the carrier frequency F_(c), sothat there is no need to adjust any communication channel on thereceiver side.

One skilled in the art will appreciate that the present invention can bepracticed by other embodiments other than those described herein, whichare provided for purposes of illustration and explanation, and not oflimitation.

The invention claimed is:
 1. An RF telemetry reception circuit for anactive implantable medical device, comprising: n antenna (104) forreceiving a double-modulation RF signal waveform, said double-modulationRF signal comprising a binary signal (a) in a baseband with a bandwidth(D_(b)) that has a first modulation by a first carrier frequency (b), togive a first modulated signal (d), and the first modulated signal (d)has a second modulation by a second carrier frequency (c), the firstcarrier frequency being a lower frequency (F_(m)) than the secondcarrier frequency (F_(c)) an input stage (108) having an inputcorresponding to said received RF signal and a bandpass filter having abandpass centered on the second carrier frequency (F_(c)), an impedancematching circuit, and an output, said output being a filtered RF signal;an envelope detector stage that is a non heterodyne detection stage,with no local oscillator and no mixer, and comprises: a first diodecircuit (120) of non-coherent detection, receiving as an input thefiltered RF signal; an active bandpass filter (122), receiving as aninput the signal (f) delivered by the first non-coherent detection diodecircuit, this active bandpass filter having a band pass centered on afrequency (2.F_(m)) that is twice that of the first frequency carrier(F_(m)) and having a bandwidth (2.D_(b)) twice that of the baseband(D_(b)), and a second diode circuit (124) of non-coherent detection,receiving as input the signal (g) delivered by the active bandpassfilter and outputting a baseband signal (h), and a digital demodulationstage (116) having an input corresponding to the baseband signal and anoutput corresponding to said binary signal of said received RF signal.2. The receiver circuit of claim 1, wherein the first diode non-coherentdetection circuit further comprises a current source (128) for diodebias.
 3. The receiver circuit of claim 1, wherein the first diodenon-coherent detection circuit further comprises a diode detector havingtwo diodes (130, 132) configured as a voltage doubler.
 4. The receivercircuit of claim 1, wherein the second carrier frequency (F_(c)) is inone of the 2.4 GHz ISM band and the 900 MHz RFID band.
 5. The receivercircuit of claim 1, wherein the first carrier frequency (F_(m)) isbetween 25 and 500 kHz.
 6. The receiver circuit of claim 1, wherein thebaseband signal (Db) has a data rate of between 5 and 100 kbps.
 7. Thereceiver circuit of claim 1, wherein the bandpass filter furthercomprises (122) a gain of between 30 and 50 dB.